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LTC1147-3.3 LTC1147-5/LTC1147L High Efficiency Step-Down Switching Regulator Controllers
FEATURES
s s s
DESCRIPTIO
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Very High Efficiency: Over 95% Possible Wide VIN Range: 3.5V* to 16V Current Mode Operation for Excellent Line and Load Transient Response High Efficiency Maintained Over Three Decades of Output Current Low 160A Standby Current at Light Loads Logic Controlled Micropower Shutdown: IQ < 20A Short-Circuit Protection Very Low Dropout Operation: 100% Duty Cycle High Efficiency in a Small Amount of Board Space Output Can Be Externally Held High in Shutdown Available in 8-Pin SO Package
The LTC(R)1147 series are step-down switching regulator controllers featuring automatic Burst ModeTM operation to maintain high efficiencies at low output currents. These devices drive an external P-channel power MOSFET at switching frequencies exceeding 400kHz using a constant off-time current mode architecture providing constant ripple current in the inductor. The operating current level is user-programmable via an external current sense resistor. Wide input supply range allows operation from 3.5V* to 14V (16V maximum). Constant off-time architecture provides low dropout regulation limited by only the RDS(ON) of the external MOSFET and resistance of the inductor and current sense resistor. The LTC1147 series incorporate automatic power saving Burst Mode operation to reduce switching losses when load currents drop below the level required for continuous operation. Standby power is reduced to only 2mW at VIN = 10V (at IOUT = 0). Load currents in Burst Mode operation are typically 0mA to 300mA. For applications where even higher efficiency is required, refer to the LTC1148 data sheet and Application Note 54.
, LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. *LTC1147L and LTC1147L-3.3 only.
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APPLICATI
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S
Notebook and Palmtop Computers Portable Instruments Battery-Operated Digital Devices Cellular Telephones DC Power Distribution Systems GPS Systems
TYPICAL APPLICATI
+
1F VIN P-DRIVE LTC1147-5 SHUTDOWN SENSE + ITH SENSE - CT GND
VIN (5.2V TO 14V)
+
P-CHANNEL Si9430DY L* 50H
CIN 100F RSENSE** 0.05
0V = NORMAL >1.5V = SHUTDOWN
VOUT 5V/2A
EFFICIENCY (%)
1000pF
RC 1k CC 3300pF
+
D1 MBRD330
CT 470pF
COUT 390F
LT1147 * F01
*COILTRONICS CTX50-2-MP **KRL SL-1-C1-0R050J
Figure 1. High Efficiency Step-Down Converter
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LTC1147-5 Efficiency
100 95 90 VIN = 10V 85 80 75 70 0.001 VIN = 6V 0.01 0.1 1
LT1147 * TA01
UO
UO
LOAD CURRENT (A)
1
LTC1147-3.3 LTC1147-5/LTC1147L ABSOLUTE AXI U RATI GS
PACKAGE/ORDER I FOR ATIO
TOP VIEW VIN 1 CT 2 ITH 3 SENSE - 4 8 7 6 5 P-DRIVE GND SHUTDOWN (VFB*) SENSE +
Input Supply Voltage (Pin 1)..................... 16V to - 0.3V Continuous Output Current (Pin 8) ...................... 50mA Sense Voltages (Pins 4, 5)........................ 10V to - 0.3V Operating Ambient Temperature Range ...... 0C to 70C Extended Commercial Temperature Range ............................... - 40C to 85C Junction Temperature (Note 1)............................ 125C Storage Temperature Range ................ - 65C to 150C Lead Temperature (Soldering, 10 sec)................. 300C
ORDER PART NUMBER LTC1147CN8-3.3 LTC1147CN8-5 LTC1147CS8-3.3 LTC1147CS8-5 LTC1147LCS8 LTC1147LCS8-3.3 S8 PART MARKING 11473 11475 1147L 1147L3
N8 PACKAGE 8-LEAD PLASTIC DIP S8 PACKAGE 8-LEAD PLASTIC SOIC * ADJUSTABLE OUTPUT VERSION
TJMAX = 125C, JA = 110C/ W (N) TJMAX = 125C, JA = 150C/ W (S)
Consult factory for Industrial and Military grade parts.
ELECTRICAL CHARACTERISTICS
SYMBOL V6 I6 VOUT PARAMETER Feedback Voltage (LTC1147L) Feedback Current (LTC1147L) Regulated Output Voltage LTC1147-3.3, LTC1147L-3.3 LTC1147-5 Output Voltage Line Regulation Output Voltage Load Regulation LTC1147-3.3, LTC1147L-3.3 LTC1147-5 Burst Mode Output Ripple IQ Input DC Supply Current (Note 2) LTC1147 Series Normal Mode Sleep Mode Sleep Mode (LTC1147-5) Shutdown LTC1147L Series Normal Mode Sleep Mode Shutdown (LTC1147L-3.3) Current Sense Threshold Voltage (Note 6) LTC1147-3.3, LTC1147L-3.3 LTC1147-5 LTC1147L
TA = 25C, VIN = 10V, VSHUTDOWN = 0V, unless otherwise noted.
MIN
q q
CONDITIONS VIN = 9V VIN = 9V ILOAD = 700mA ILOAD = 700mA VIN = 7V to 12V, ILOAD = 50mA 5mA < ILOAD < 2A 5mA < ILOAD < 2A ILOAD = 0A (Note 5) 4V < VIN < 12V 4V < VIN < 12V 5V < VIN < 12V VSHUTDOWN = 2.1V, 4V < VIN < 12V 3.5V < VIN < 12V 3.5V < VIN < 12V VSHUTDOWN = 2.1V, 3.5V < VIN < 12V VSENSE- = VOUT + 100mV (Forced) VSENSE- = VOUT - 100mV (Forced) VSENSE- = VOUT + 100mV (Forced) VSENSE- = VOUT - 100mV (Forced) VSENSE- = 5V, V6 = VOUT/4 + 25mV (Forced) VSENSE- = 5V, V6 = VOUT/4 - 25mV (Forced)
q q
TYP 1.25 0.2
MAX 1.29 1 3.43 5.20 40 65 100
UNITS V A V V mV mV mV mVP-P
1.21
q q
3.23 4.90 - 40
3.33 5.05 0 40 60 50
VOUT
1.6 160 160 10 1.6 160 10 25 150 25 150 25 150
2.1 230 230 20 2.1 230 20
V5 - V 4
130 130 130
170 170 170
2
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mA A A A mA A A mV mV mV mV mV mV
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LTC1147-3.3 LTC1147-5/LTC1147L
ELECTRICAL CHARACTERISTICS
SYMBOL V6 I6 I2 tOFF tr, tf PARAMETER Shutdown Pin Threshold LTC1147-3.3, LTC1147-5, LTC1147L-3.3 Shutdown Pin Input Current LTC1147-3.3, LTC1147-5, LTC1147L-3.3, CT Pin Discharge Current Off-Time (Note 3) Driver Output Transition Times
TA = 25C, VIN = 10V, VSHUTDOWN = 0V, unless otherwise noted.
MIN 0.6 TYP 0.8 1.2 50
q
CONDITIONS
MAX 2 5 90 10 6 200
UNITS V A A A s ns
0V < VSHUTDOWN < 8V, VIN = 16V VOUT in Regulation, VSENSE - = VOUT VOUT = 0V CT = 390pF, ILOAD = 700mA CL = 3000pF (Pins 8), VIN = 6V
70 2 5 100
4
- 40C TA 85C (Note 4), VIN = 10V, unless otherwise noted.
SYMBOL V6 VOUT PARAMETER Feedback Voltage (LTC1147L) Regulated Output Voltage LTC1147-3.3, LTC1147L-3.3 LTC1147-5 Input DC Supply Current (Note 2) LTC1147 Series Normal Mode Sleep Mode Sleep Mode (LTC1147-5) Shutdown LTC1147L Series Normal Mode Sleep Mode Shutdown (LTC1147L-3.3) Current Sense Threshold Voltage (Note 6) LTC1147-3.3 LTC1147-5 LTC1147L V6 tOFF Shutdown Pin Threshold LTC1147-3.3, LTC1147-5, LTC1147L-3.3, Off-Time (Note 3) CONDITIONS VIN = 9V VIN = 9V ILOAD = 700mA ILOAD = 700mA (Note 5) 4V < VIN < 12V 4V < VIN < 12V 5V < VIN < 12V VSHUTDOWN = 2.1V, 4V < VIN < 12V 3.5V < VIN < 12V 3.5V < VIN < 12V VSHUTDOWN = 2.1V, 3.5V < VIN <12V VSENSE - = VOUT + 100mV (Forced) VSENSE - = VOUT - 100mV (Forced) VSENSE - = VOUT + 100mV (Forced) VSENSE - = VOUT - 100mV (Forced) VSENSE- = 5V, 6V = VOUT/4 + 25mV (Forced) VSENSE- = 5V, 6V = VOUT/4 - 25mV (Forced) 0V < VSHUTDOWN < 8V, VIN = 16V CT = 390pF, ILOAD = 700mA
q q q q
MIN 1.20 3.17 4.85
TYP 1.25 3.33 5.05
MAX 1.30 3.4 5.2
UNITS V V V
IQ
1.6 160 160 10 1.6 160 10 25 150 25 150 25 150 0.8 5
2.4 260 260 22 2.4 260 22
mA A A A mA A A mV mV mV mV mV mV V s
V5 - V4
q q
125 125 125 0.55 3.8
175 175 175 2 6.5
The q denotes specifications which apply over the full operating temperature range. Note 1: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC1147CN8-3.3, LTC1147CN8-5: TJ = TA + (PD x 110C/W) LTC1147LCS, LTC1147CS8-3.3, LTC1147CS8-5: TJ = TA + (PD x 150C/W) Note 2: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information.
Note 3: In applications where RSENSE is placed at ground potential, the offtime increases approximately 40%. Note 4: The LTC1147-3.3/LTC1147-5 are not tested and not quality assurance sampled at - 40C and 85C. These specifications are guaranteed by design and/or correlation. Note 5: The LTC1147L and LTC1147L-3.3 allow operation to VIN = 3.5V Note 6: The LTC1147L is tested with external feedback resistors resulting in a nominal output voltage of 2.5V.
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LTC1147-3.3 LTC1147-5/LTC1147L
TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Input Voltage
100 98 96 FIGURE 1 CIRCUIT
EFFICIENCY (%)
94 92 90 88 86 84 82 80 0 4
VOUT (mV)
VOUT (mV)
ILOAD = 1A
ILOAD = 100mA
12 8 INPUT VOLTAGE (V)
LTC1147 * TPC01
DC Supply Current
2.1 1.8
SUPPLY CURRENT (mA)
NOT INCLUDING GATE CHARGE CURRENT SUPPLY CURRENT (A)
NORMALIZED FREQUENCY
1.5 1.2 0.9 0.6 0.3 0 0 2 4
ACTIVE MODE
SLEEP MODE 6 8 10 12 14 INPUT VOLTAGE (V) 16 18
LTC1147 * TPC04
Gate Charge Supply Current
14 12 10
80 70
GATE CHARGE CURRENT (mA)
SENSE VOLTAGE (mV)
OFF-TIME (s)
8 6 4
QP = 50nC
QP = 29nC 2 0 20 260 80 200 140 OPERATING FREQUENCY (kHz)
LTC1147 * TPC07
4
UW
16
Line Regulation
40 30 20 10 0 -10 -20 -30 -40 0 4 8 12 16
LTC1147 * TPC02
Load Regulation
20 FIGURE 1 CIRCUIT RSENSE = 0.05
FIGURE 1 CIRCUIT ILOAD = 1A
0 -20 -40 -60 -80 -100 0 0.5 VIN = 12V
VIN = 6V
INPUT VOLTAGE (V)
1 1.5 LOAD CURRENT (A)
2
2.5
LTC1147 * TPC03
Supply Current in Shutdown
20 18 16 14 12 10 8 6 4 2 0 0 2 4 6 8 10 12 14 INPUT VOLTAGE (V) 16 18 VSHUTDOWN = 2V (NOT AVAILABLE ON LTC1147L)
1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0
Operating Frequency vs (VIN - VOUT)
VOUT = 5V 0C 70C 25C
0
2
4 8 10 6 (VIN - VOUT) VOLTAGE (V)
12
LTC1147 * TPC05
LTC1148 * TPC06
Off-Time vs VOUT
VSENSE - = VOUT
Current Sense Threshold Voltage
175 150 125 100 75 50 25 0
3 4 5
MAXIMUM THRESHOLD
60 50 40 30 20 10 LTC1147-3.3 0 0 1 2 LTC1147-5
MINIMUM THRESHOLD
0
20
OUTPUT VOLTAGE (V)
LTC1147 * TPC08
60 40 TEMPERATURE (C)
80
100
LTC1147 * TPC09
LTC1147-3.3 LTC1147-5/LTC1147L
PI FU CTIO S
Pin 1 (VIN): Main Supply Pin. Must be closely decoupled to ground pin 7. Pin 2 (CT): External capacitor CT from pin 2 to ground sets the operating frequency. The actual frequency is also dependent upon the input voltage. Pin 3 (ITH): Gain Amplifier Decoupling Point. The current comparator threshold increases with the pin 3 voltage. Pin 4 (Sense -): Connects to internal resistive divider which sets the output voltage. Pin 4 is also the (-) input for the current comparator. Pin 5 (Sense +): The (+) input to the current comparator. A built-in offset between pins 4 and 5 in conjunction with RSENSE sets the current trip threshold. Pin 6 (Shutdown/VFB): When grounded, the fixed output versions of the LTC1147 family operate normally. Pulling pin 6 high holds the P-channel MOSFET off and puts the LTC1147 in micropower shutdown mode. Requires CMOS logic signal with tr, tf < 1s. Do not leave this pin floating. On the LTC1147L this pin serves as the feedback pin from an external resistive divider used to set the output voltage. Pin 7 (GND): Two independent ground lines must be routed separately to: 1) the (-) terminal of COUT, and 2) the cathode of the Schottky diode and (-) terminal of CIN. Pin 8 (P-Drive): High current drive for the P-channel MOSFET. Voltage swing at this pin is from VIN to ground.
FU CTIO AL DIAGRA
V SLEEP
R Q S
C
25mV TO 150mV
-
+
5pF VOS
+
S
T
13k ITH 3 G
2 CT
OFF-TIME CONTROL
VIN SENSE - SHUTDOWN 6 REFERENCE
LTC1147 * FD
-
-
VTH2
VTH1
-
+
+
-
+
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+
-
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Pin 6 Connection Shown For LTC1147-3.3 and LTC1147-5; Changes Create LTC1147L.
SENSE+ 1 VIN 5 VFB 6 7 GROUND SENSE - 4
8 P-DRIVE
1.25V 100k
5
LTC1147-3.3 LTC1147-5/LTC1147L
OPERATIO
The LTC1147 series use a current mode, constant off-time architecture to switch an external P-channel power MOSFET. Operating frequency is set by an external capacitor at CT (pin 2). The output voltage is sensed by an internal voltage divider connected to Sense- (pin 4). A voltage comparator V, and a gain block G, compare the divided output voltage with a reference voltage of 1.25V. To optimize efficiency, the LTC1147 series automatically switch between two modes of operation, burst and continuous. The voltage comparator is the primary control element when the device is in Burst Mode operation, while the gain block controls the output voltage in continuous mode. During the switch "on" cycle in continuous mode, current comparator C monitors the voltage between pins 4 and 5 connected across an external shunt in series with the inductor. When the voltage across the shunt reaches its threshold value, the P-drive output is switched to VIN, turning off the P-channel MOSFET. The timing capacitor connected to pin 2 is now allowed to discharge at a rate determined by the off-time controller. The discharge current is made proportional to the output voltage (measured by pin 4) to model the inductor current, which decays at a rate which is also proportional to the output voltage. When the voltage on the timing capacitor has discharged past VTH1, comparator T trips, setting the flip-flop. This causes the P-drive output to go low turning the P-channel MOSFET back on. The cycle then repeats. As the load current increases, the output voltage decreases slightly. This causes the output of the gain stage
APPLICATIO S I FOR ATIO
LTC1147L Adjustable Applications
When an output voltage other than 3.3V or 5V is required, the LTC1147L adjustable version is used with an external resistive divider from VOUT to VFB pin 6 (See Figure 7). The regulated voltage is determined by:
VOUT = 1.25 1 +
)
R2 R1
)
6
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(Refer to Functional Diagram)
(pin 3) to increase the current comparator threshold, thus tracking the load current. The sequence of events for Burst Mode operation is very similar to continuous operation with the cycle interrupted by the voltage comparator. When the output voltage is at or above the desired regulated value, the P-channel MOSFET is held off by comparator V and the timing capacitor continues to discharge below VTH1. When the timing capacitor discharges past VTH2, voltage comparator S trips, causing the internal sleep line to go low. The circuit now enters sleep mode with the power MOSFET turned off. In sleep mode, a majority of the circuitry is turned off, dropping the quiescent current from 1.6mA to 160A. The load current is now being supplied from the output capacitor. When the output voltage has dropped by the amount of hysteresis in comparator V, the P-channel MOSFET is again turned on and this process repeats. To avoid the operation of the current loop interfering with Burst Mode operation, a built-in offset VOS is incorporated in the gain stage. This prevents the current comparator threshold from increasing until the output voltage has dropped below a minimum threshold. Using constant off-time architecture, the operating frequency is a function of the input voltage. To minimize the frequency variation as dropout is approached, the off-time controller increases the discharge current as VIN drops below VOUT + 1.5V. In dropout the P-channel MOSFET is turned on continuously (100% duty cycle), providing low dropout operation with VOUT VIN.
To prevent stray pickup a 100pF capacitor is suggested across R1 located close to the LTC1147L. For Figure 1 applications with VOUT below 2V, or when RSENSE is moved to ground, the current sense comparator inputs operate near ground. When the current comparator is operated at less than 2V common mode, the off-time increases approximately 40%, requiring the use of a smaller timing capacitor CT.
LTC1147-3.3 LTC1147-5/LTC1147L
APPLICATIO S I FOR ATIO
The basic LTC1147 application circuit is shown in Figure 1. External component selection is driven by the load requirement, and begins with the selection of RSENSE. Once RSENSE is known, CT and L can be chosen. Next, the power MOSFET and D1 are selected. Finally, CIN and COUT are selected and the loop is compensated. The circuit shown in Figure 1 can be configured for operation up to an input voltage of 16V. If the application requires higher input voltage, then the synchronous switched LTC1149 should be used. Consult factory for lower minimum input voltage version. RSENSE Selection for Output Current RSENSE is chosen based on the required output current. The LTC1147 series current comparator has a threshold range which extends from a minimum of 25mV/ RSENSE to a maximum of 150mV/RSENSE. The current comparator threshold sets the peak of the inductor ripple current, yielding a maximum output current IMAX equal to the peak value less half the peak-to-peak ripple current. For proper Burst Mode operation, IRIPPLE(P-P) must be less than or equal to the minimum current comparator threshold. Since efficiency generally increases with ripple current, the maximum allowable ripple current is assumed, i.e., IRIPPLE(P-P) = 25mV/RSENSE (See CT and L Selection for Operating Frequency). Solving for RSENSE and allowing a margin for variations in the LTC1147 series and external component values yields:
RSENSE = 100mV IMAX
RSENSE ()
A graph for selecting RSENSE versus maximum output current is given in Figure 2. The load current below in which Burst Mode operation commences, IBURST and the peak short-circuit current ISC(PK), both track IMAX. Once RSENSE has been chosen, IBURST and ISC(PK) can be predicted from the following:
IBURST 15mV RSENSE ISC(PK) = 150mV RSENSE
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0.20 0.15 0.10 0.05 0 0 1 3 4 2 MAXIMUM OUTPUT CURRENT (A) 5
LTC1147 * F02
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Figure 2. Selecting RSENSE
The LTC1147 series automatically extend tOFF during a short circuit to allow sufficient time for the inductor current to decay between switch cycles. The resulting ripple current causes the average short-circuit current ISC(AVG) to be reduced to approximately IMAX. L and CT Selection for Operating Frequency The LTC1147 series use a constant off-time architecture with tOFF determined by an external timing capacitor CT. Each time the P-channel MOSFET switch turns on, the voltage on CT is reset to approximately 3.3V. During the off-time, CT is discharged by a current which is proportional to VOUT. The voltage on CT is analogous to the current in inductor L, which likewise decays at a rate proportional to VOUT. Thus the inductor value must track the timing capacitor value. The value of CT is calculated from the desired continuous mode operating frequency:
CT = VIN - VOUT 1 4xf V +V 1.3 x 10 IN D
)
)
Where VD is the drop across the Schottky diode. A graph for selecting CT versus frequency including the effects of input voltage is given in Figure 3.
7
LTC1147-3.3 LTC1147-5/LTC1147L
APPLICATIO S I FOR ATIO
1000 VSENSE - = VOUT = 5V 800
CAPACITANCE (pF)
600
400 VIN = 12V 200 VIN = 7V VIN = 10V
0 0 200 100 FREQUENCY (kHz) 300
LTC1147 * F03
Figure 3. Timing Capacitor Value
As the operating frequency is increased the gate charge losses will reduce efficiency (see Efficiency Considerations). The complete expression for operating frequency is given by:
f
where:
1 tOFF
)
1-
VOUT VIN
)
tOFF = 1.3 x 104 x CT x
))
VREG VOUT
VREG is the desired output voltage (i.e., 5V, 3.3V). VOUT is the measured output voltage. Thus VREG/VOUT = 1 in regulation. Note that as VIN decreases, the frequency decreases. When the input to output voltage differential drops below 1.5V, the LTC1147 reduces tOFF by increasing the discharge current in CT. This prevents audible operation prior to dropout. Once the frequency has been set by CT, the inductor L must be chosen to provide no more than 25mV/RSENSE of peak-to-peak inductor ripple current. This results in a minimum required inductor value of: LMIN = 5.1 x 105 x RSENSE x CT x VREG As the inductor value is increased from the minimum value, the ESR requirements for the output capacitor are eased at the expense of efficiency. If too small an
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inductor is used, the inductor current will become discontinuous before the LTC1147 series enters Burst Mode operation. A consequence of this is that the LTC1147 series will delay entering Burst Mode operation and efficiency will be degraded at low currents. Inductor Core Selection Once the minimum value for L is known, the type of inductor must be selected. Highest efficiency will be obtained using ferrite, Kool M(R) (from Magnetics, Inc.) or molypermalloy (MPP) cores. Lower cost powdered iron cores provide suitable performance but cut efficiency by 3% to 5%. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple which can cause Burst Mode operation to be falsely triggered in the LTC1147. Do not allow the core to saturate! Kool M is a very good, low loss core material for toroids with a "soft" saturation characteristic. Molypermalloy is slightly more efficient at high (>200kHz) switching frequencies but quite a bit more expensive. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, new designs for surface mount are available from Coiltronics, Sumida and Beckman Industrial Corp. which do not increase the height significantly. Power MOSFET Selection An external P-channel power MOSFET must be selected for use with the LTC1147 series. The main selection criteria for the power MOSFET are the threshold voltage VGS(TH) and "on" resistance RDS(ON).
Kool M is a registered trademark of Magnetics, Inc.
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LTC1147-3.3 LTC1147-5/LTC1147L
APPLICATIO S I FOR ATIO
The minimum input voltage determines whether a standard threshold or logic-level threshold MOSFET must be used. For VIN > 8V, a standard threshold MOSFET (VGS(TH) < 4V) may be used. If VIN is expected to drop below 8V, a logic-level threshold MOSFET (VGS(TH) < 2.5V) is strongly recommended. When a logic-level MOSFET is used, the LTC1147 supply voltage must be less than the absolute maximum VGS ratings for the MOSFET. The maximum output current IMAX determines the RDS(ON) requirement for the power MOSFET. When the LTC1147 series is operating in continuous mode, the simplifying assumption can be made that either the MOSFET or Schottky diode is always conducting the average load current. The duty cycles for the MOSFET and diode are given by: V P-Ch Duty Cycle = OUT VIN Schottky Diode Duty Cycle = (VIN - VOUT + VD) VIN
From the duty cycle the required RDS(ON) for the MOSFET can be derived:
P-Ch RDS(ON) = VIN x PP VOUT x IMAX2 x (1 + P)
where PP is the allowable power dissipation and P is the temperature dependency of RDS(ON). PP will be determined by efficiency and/or thermal requirements (see Efficiency Considerations). (1 + ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but = 0.007/C can be used as an approximation for low voltage MOSFETs. Output Diode Selection (D1) The Schottky diode D1 shown in Figure 1 only conducts during the off-time. It is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. The most stressful condition for the output diode is under short circuit (VOUT = 0). Under this condition the diode must safely handle ISC(PK) at close to 100% duty cycle. Under normal load conditions the average current conducted by the diode is:
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ID1 = (VIN - VOUT + VD) (ILOAD) VIN
Remember to keep lead lengths short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. The forward voltage drop allowable in the diode is calculated from the maximum short-circuit current as:
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VF
PD ISC(PK)
where PD is the allowable power dissipation and will be determined by efficiency and/or thermal requirements (see Efficiency Considerations). CIN and COUT Selection In continuous mode, the source current of the P-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by:
CIN Required IRMS IMAX
[VOUT(VIN - VOUT)]1/2 VIN
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. An additional 0.1F to 1F ceramic decoupling capacitor is also required on VIN (pin 1) for high frequency decoupling. The selection of COUT is driven by the required effective series resistance (ESR). The ESR of COUT must be less than twice the value of RSENSE for proper operation of the LTC1147 : COUT Required ESR < 2RSENSE
9
LTC1147-3.3 LTC1147-5/LTC1147L
APPLICATIO S I FOR ATIO
Optimum efficiency is obtained by making the ESR equal to RSENSE. As the ESR is increased up to 2RSENSE, the efficiency degrades by less than 1%. If the ESR is greater than 2RSENSE, the voltage ripple on the output capacitor will prematurely trigger Burst Mode operation, resulting in disruption of continuous mode and an efficiency hit which can be several percent. Manufacturers such as Nichicon and United Chemicon should be considered for high performance capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR/size ratio of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications multiple capacitors may have to be paralleled to meet the capacitance, ESR, or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. For example, if 200F/10V is called for in an application requiring 3mm height, two AVX 100F/10V (P/N TPSD 107K010) could be used. Consult the manufacturer for other specific recommendations. At low supply voltages, a minimum capacitance at COUT is needed to prevent an abnormal low frequency operating
1000 L = 50H RSENSE = 0.02 800 L = 25H RSENSE = 0.02
COUT (F)
600
400
200 L = 50H RSENSE = 0.05 0 0 1 3 4 2 (VIN - VOUT) VOLTAGE (V) 5
LTC1147 * F04
Figure 4. Minimum Value of COUT
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mode (see Figure 4). When COUT is made too small, the output ripple at low frequencies will be large enough to trip the voltage comparator. This causes Burst Mode operation to be activated when the LTC1147 series would normally be in continuous operation. The effect is most pronounced with low values of RSENSE and can be improved by operating at higher frequencies with lower values of L. The output remains in regulation at all times. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in DC (resistive) load current. When a load step occurs, VOUT shifts by an amount equal to ILOAD x ESR, where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT until the regulator loop adapts to the current change and returns VOUT to its steady state value. During this recovery time VOUT can be monitored for overshoot or ringing which would indicate a stability problem. The external components shown in the Figure 1 circuit will prove adequate compensation for most applications. A second, more severe transient is caused by switching in loads with large (>1F) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately 25 x CLOAD. Thus a 10F capacitor would require a 250s rise time, limiting the charging current to about 200mA. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% - (L1 + L2 + L3 + ...)
W
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LTC1147-3.3 LTC1147-5/LTC1147L
APPLICATIO S I FOR ATIO
where L1, L2, etc., are the individual losses as a percentage of input power. (For high efficiency circuits only small errors are incurred by expressing losses as a percentage of output power). Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1147 circuits: 1) LTC1147 DC bias current, 2) MOSFET gate charge current, 3) I2R losses, and 4) voltage drop of the Schottky diode. 1. The DC supply current is the current which flows into VIN (pin 1) less the gate charge current. For VIN = 10V the LTC1147 series DC supply current is 160A for no load, and increases proportionally with load up to a constant 1.6mA after the LTC1147 series has entered continuous mode. Because the DC bias current is drawn from VIN, the resulting loss increases with input voltage. For VIN = 10V the DC bias losses are generally less than 1% for load currents over 30mA. However, at very low load currents the DC bias current accounts for nearly all of the loss. 2. MOSFET gate charge current results from switching the gate capacitance of the power MOSFET. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN which is typically much larger than the DC supply current. In continuous mode, IGATECHG = f(QP). The typical gate charge for a 0.135 P-channel power MOSFET is 40nC. This results in IGATECHG = 4mA in 100kHz continuous operation for a 2% to 3% typical mid-current loss with VIN = 10V. Note that the gate charge loss increases directly with both input voltage and operating frequency. This is the principal reason why the highest efficiency circuits operate at moderate frequencies. Furthermore, it argues against using a larger MOSFET than necessary to control I2R losses, since overkill can cost efficiency as well as money! 3. I2R losses are easily predicted from the DC resistances of the MOSFET, inductor, and current shunt. In continuous mode the average output current flows through L and RSENSE, but is "chopped" between the
EFFICIENCY/LOSS (%)
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P-channel and Schottky diode. The MOSFET RDS(ON) multiplied by the P-channel duty cycle can be summed with the resistances of L and RSENSE to obtain I2R losses. For example, if RDS(ON) = 0.1, RL = 0.15, and RSENSE = 0.05, then the total resistance is 0.3 at VIN 2VOUT. This results in losses ranging from 3% to 10% as the output current increases from 0.5A to 2A. I2R losses cause the efficiency to roll off at high output currents. 4. The Schottky diode is a major source of power loss at high currents and gets worse at high input voltages. The diode loss is calculated by multiplying the forward voltage drop times the Schottky diode duty cycle multiplied by the load current. For example, assuming a duty cycle of 50% with a Schottky diode forward voltage drop of 0.4V, the loss increases from 0.5% to 8% as the load current increases from 0.5A to 2A. Figure 5 shows how the efficiency losses in a typical LTC1147 series regulator end up being apportioned. The gate charge loss is responsible for the majority of the efficiency lost in the mid-current region. If Burst Mode operation was not employed at low currents, the gate charge loss alone would cause efficiency to drop to unacceptable levels. With Burst Mode operation, the DC supply current represents the lone (and unavoidable) loss component which continues to become a higher percentage as output current is reduced. As expected, the I2R losses and Schottky diode loss dominate at high load currents.
100 GATE CHARGE 95 LTC1147 IQ 90 SCHOTTKY DIODE I2R 85 80 0.01 0.03 0.3 1 0.1 OUTPUT CURRENT (A) 3
LTC1147 * F05
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Figure 5. Efficiency Loss
11
LTC1147-3.3 LTC1147-5/LTC1147L
APPLICATIO S I FOR ATIO
Other losses including CIN and COUT ESR dissipative losses, MOSFET switching losses, and inductor core losses, generally account for less than 2% total additional loss. Design Example As a design example, assume VIN = 5V (nominal), VOUT = 3.3V, IMAX = 1A, and f = 130kHz; RSENSE, CT and L can immediately be calculated: RSENSE = 100mV/1A = 0.1 tOFF = (1/130kHz) x [1 - (3.3/5)] = 2.61s CT = 2.61s/(1.3 x 104) = 220pF L = 5.1 x 105 x 0.1 x 220pF x 3.3V = 33H Assume that the MOSFET dissipation is to be limited to PP = 250mW. If TA = 50C and the thermal resistance of the MOSFET is 50C/ W, then the junction temperatures will be 63C and P = 0.007(63 - 25) = 0.27. The required RDS(ON) for the MOSFET can now be calculated:
P-Ch RDS(ON) = 5(0.25) = 0.3 3.3(1)2 (1.27)
The P-channel requirement can be met by a Si9430DY. Note that the most stringent requirement for the Schottky diode is with VOUT = 0 (i.e., short circuit). During a continuous short circuit, the worst-case Schottky diode dissipation rises to:
PD = ISC(AVG) x VD
3.3V
With the 0.1 sense resistor ISC(AVG) = 1A will result, increasing the 0.4V Schottky diode dissipation to 0.4W. CIN will require an RMS current rating of at least 0.5A at temperature, and COUT will require an ESR of 0.1 for optimum efficiency. Now allow VIN to drop to its minimum value. At lower input voltages the operating frequency will decrease and the P-channel will be conducting most of the time, causing the power dissipation to increase. At VIN(MIN) = 4.5V, the frequency will decrease and the P-channel will be conducting most of the time causing its power dissipation to increase. At VIN(MIN) = 4.5V:
(a) CONTINUOUS MODE OPERATION
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fMIN = 1 3.3 = 102kHz 1- 2.61s 4.5
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)
)
PP =
3.3(0.125)(1A)2(1.27) = 116mW 4.5
This last step is necessary to assure that the power dissipation and junction temperature of the P-channel are not exceeded. Troubleshooting Hints Since efficiency is critical to LTC1147 series applications, it is very important to verify that the circuit is functioning correctly in both continuous and Burst Mode operation. The waveform to monitor is the voltage on the timing capacitor pin 2. In continuous mode (ILOAD > IBURST) the voltage on the CT pin should be a sawtooth with a 0.9VP-P swing. This voltage should never dip below 2V as shown in Figure 6a. When load currents are low (ILOAD < IBURST) Burst Mode operation occurs. The voltage on the CT pin now falls to ground for periods of time as shown in Figure 6b. During this time the LTC1147 series are in sleep mode with the quiescent current reduced to 160A. The inductor current should also be monitored. Look to verify that the peak-to-peak ripple current in continuous mode operation is approximately the same as in Burst Mode operation.
0V
3.3V
0V (b) Burst Mode OPERATION
LTC1147 * F06
Figure 6. CT Waveforms
If pin 2 is observed falling to ground at high output currents, it indicates poor decoupling or improper grounding. Refer to the Board Layout Checklist.
LTC1147-3.3 LTC1147-5/LTC1147L
APPLICATIO S I FOR ATIO
Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1147 series. These items are also illustrated graphically in the layout diagram of Figure 7. Check the following in your layout: 1. Are the signal and power grounds segregated? The LTC1147 ground (pin 7) must return separately to a) the power and b) signal grounds. The power ground (a) returns to the source anode of the Schottky diode, and (-) plate of CIN, which should have lead lengths as short as possible. The signal ground (b) connects to the (-) plate of COUT. 2. Does the LTC1147 Sense - (pin 4) connect to a point close to RSENSE and the (+) plate of COUT?
+
VIN
BOLD LINES INDICATE HIGH CURRENT PATHS
+
CIN
-
1F
LTC1147-3.3 LTC1147-5 (LTC1147L) 1 2 3 VIN CT ITH SENSE - P-DRIVE GROUND 8 7 6 5 SHUTDOWN
390pF
3300pF 1k
SHUTDOWN (VFB)
SENSE +
4
1000pF
Figure 7. LTC1147 Layout Diagram (See Board Layout Checklist)
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3. Are the Sense - and Sense + leads routed together with minimum PC trace spacing? The 1000pF capacitor between pins 4 and 5 should be as close as possible to the LTC1147. 4. Does the (+) plate of CIN connect to the source of the P-channel MOSFET as closely as possible? This capacitor provides the AC current to the P-channel MOSFET. 5. Is the input decoupling capacitor (0.1F/1F) connected closely between VIN (pin 1) and ground (pin 7)? This capacitor carries the MOSFET driver peak currents. 6. On fixed output versions, is the Shutdown (pin 6) actively pulled to ground during normal operation? The Shutdown pin is high impedance and must not be allowed to float.
P-CH L RSENSE D1
+
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+
COUT
+
VOUT
-
R1 100pF
R2
OUTPUT DIVIDER REQUIRED WITH ADJUSTABLE VERSION ONLY
LTC1147 * F07
For additional High Efficiency application circuits see Application Note 54.
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LTC1147-3.3 LTC1147-5/LTC1147L
TYPICAL APPLICATIO S
VIN 3.5V TO 12V
1 2 3
CT 120pF
CC 3300pF RC 1k
4
0.01F
RSENSE** 0.068
*SUMIDA CDR74B-100LC **IRC LRC-LR2010-01-R068-F
Figure 8. 3.3V Low Dropout High Efficiency Regulator
VIN 10V TO 14V
Si9400DY 0.1F
+
D1 MBRS130LT3
CIN 33F 25V
LTC1147L 1 2 VIN CT ITH SENSE - P-DRIVE GROUND SHUTDOWN (VFB) SENSE + 8 7 6 5
L* 220H
1.0
0.8
3
CC 3300pF 4 RC 1k
0.01F
RSENSE** 0.22
R2 6.8k
COUT 100F 16V AVX
IOUT (A)
CT 300pF
100pF
R1 1.2k
0.6
0.4
0.2
VOUT
0
*SUMIDA CDR74-221 **IRC LRC-LR2010-01-R068-F
LTC1147 * F09
Figure 9. Precision Constant Current Source
Figure 10. IOUT vs. VOUT For Figure 9.
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+
+
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VIN CT ITH
+
Si9433DY D1 MBRS130LT3
CIN 47F 16V
0.1F L* 10H LTC1147L-3.3 P-DRIVE GROUND SHUTDOWN SENSE + 8 7 6 5 SHUTDOWN COUT 100F 10V AVX
SENSE -
VOUT 3.3V/1.25A
LTC1147 * F08
VIN = 12.6V
0
2
6 4 VOUT (V)
8
10
LTC1147 F10
LTC1147-3.3 LTC1147-5/LTC1147L
TYPICAL APPLICATIO S
VIN 3.5V TO 12V
1 2 3
CT 120pF
CC 3300pF RC 1k
4
0.01F
RSENSE** 0.05
*COILTRONICS CTX10-4 **IRC LR2512-01-0R050-G
Figure 11. 2.5V, 2A Regulator
Si9430DY
D1 MBRS130LT3
0.1F L* 20H LTC1147-3.3 1 2 3 VIN CT ITH SENSE - P-DRIVE GROUND SHUTDOWN SENSE + 8 7 6 5 SHUTDOWN COUT 220F 10V AVX
CT 220pF
CC 3300pF RC 1k
4
0.01F
RSENSE** 0.05
*COILTRONICS CTX20-4 **KRL SP-1/2-A1-OR050
Figure 12. 3.3V, 2A Output High Efficiency Regulator
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
+
+
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CT
+
Si9433DY D1 MBRD330
CIN 15F 25V x 2
0.1F L* 10H LTC1147L VIN P-DRIVE GROUND SHUTDOWN (VFB) SENSE + 8 7 6 5 R2 49.9k 1% I00pF R1 49.9k 1% COUT 220F 10V x 2 AVX
ITH SENSE -
2.5V/2A
LTC1147 * F11
VIN 4V TO 14V
+
CIN 22F 25V x 2
LTC1147 * F12
VOUT 3.3V/2A
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LTC1147-3.3 LTC1147-5/LTC1147L
PACKAGE DESCRIPTIO U
Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead Plastic DIP
0.400 (10.160) MAX 8 7 6 5
0.250 0.010 (6.350 0.254)
1
2
3
4
0.300 - 0.320 (7.620 - 8.128)
0.045 - 0.065 (1.143 - 1.651)
0.130 0.005 (3.302 0.127)
0.009 - 0.015 (0.229 - 0.381)
0.065 (1.651) TYP 0.125 (3.175) MIN 0.020 (0.508) MIN
(
+0.025 0.325 -0.015 8.255 +0.635 -0.381
)
0.045 0.015 (1.143 0.381) 0.100 0.010 (2.540 0.254)
0.018 0.003 (0.457 0.076)
N8 0393
S8 Package 8-Lead Plastic SOIC
0.189 - 0.197 (4.801 - 5.004) 8 7 6 5
0.228 - 0.244 (5.791 - 6.197)
0.150 - 0.157 (3.810 - 3.988)
1 0.010 - 0.020 x 45 (0.254 - 0.508) 0.008 - 0.010 (0.203 - 0.254) 0.016 - 0.050 0.406 - 1.270 0- 8 TYP
2
3
4
0.053 - 0.069 (1.346 - 1.752) 0.004 - 0.010 (0.101 - 0.254)
0.014 - 0.019 (0.355 - 0.483)
0.050 (1.270) BSC
SO8 0393
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Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7487
(408) 432-1900 q FAX: (408) 434-0507 q TELEX: 499-3977
LT/GP 0195 5K REV B * PRINTED IN USA
(c) LINEAR TECHNOLOGY CORPORATION 1993

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